This invention relates to electronic instruments for detecting and/or measuring electrical signals and, more particularly, to electronic instruments for detecting and/or measuring the frequency spectrum, e.g., DC through radio-frequency (RF), of electrical signals. Specifically, one embodiment of the invention provides a routing yttrium-iron-garnet (YIG) tuned mixer particularly adaptable for use in an electronic instrument known as a spectrum analyzer. Such a mixer, which can be an odd harmonic mixer or even harmonic mixer, but is preferably a switched odd or even harmonic mixer, can also be combined with at least one additional YIG-tuned resonator to form a routing YIG-tuned resonator filter and mixer.
Generally, a spectrum analyzer is a scanning receiver that displays power and modulation characteristics of electrical input signals over a selected frequency band. To cover a broad frequency range, e.g., from 0 to 26.5 GHz, an input signal is preferably divided into low-frequency and high-frequency portions. One aspect of the invention is concerned with effectively routing the input signal between the low- and high-frequency signal processing sections of the spectrum analyzer, depending upon the frequency of the input signal.
In this regard, FIG. 1 shows a block diagram illustrating the initial signal processing circuit, or "front end," of a conventional spectrum analyzer. Initially, all input signals, e.g., signals in the frequency range from 0 to 26.5 GHz, applied to an input 11 of the spectrum analyzer pass through a step attenuator 12 and are directed to a mechanical microwave relay switch 13 which selectively routes the input signals to the low-frequency signal processing section on line 14 or to the high-frequency signal processing section on line 15. Unfortunately, mechanical microwave relay switches are slow and wear out under heavy use.
On the one hand, low-frequency input signals, e.g., input signals having a frequency under 2.9 GHz, are applied to a low-frequency signal processing section of the spectrum analyzer. Input signals from 0 to 2.9 GHz are switched to a first converter 16 comprising a lowpass filter 17, a swept, YIG-tuned local oscillator (LO) 18, a mixer 19, and a bandpass filter 20 which up-convert the low-frequency input signals to a fixed intermediate frequency (IF) output signal that is applied to a low-frequency analysis circuit (not shown) of the spectrum analyzer.
On the other hand, high-frequency signals, e.g., RF (microwave) input signals having a frequency between 2.7 and 26.5 GHz, are passed through a tunable bandpass filter 21. The passed RF input signal is then down-converted by a harmonic mixer 22. The harmonic mixer 22 combines the RF input signal with a signal produced by a local oscillator (LO) 23, or a harmonic of the signal from the LO, to produce a predetermined IF output signal at a frequency suitable for further processing by the high-frequency analysis circuit (not shown) of the spectrum analyzer.
A spectrum analysis measurement is performed on a high-frequency RF input signal by sweeping the LO signal frequency over the frequency range of interest, while the predetermined IF frequency is monitored. The graph in FIG. 2 illustrates the result of the down conversion by the harmonic mixer 22 shown in FIG. 1, evidencing the relationship between the LO, RF, and predetermined IF frequencies. In FIG. 2, the vertical axis represents signal power, and the horizontal axis represents signal frequency. The predetermined IF signal 25 has a frequency equal to the difference between the LO signal (or harmonic) 27 and the RF input signal 29, so that the RF input signal is measured by monitoring a set IF frequency, below the LO signal frequency, at f.sub.RF =(n)f.sub.LO -f.sub.IF. However, an image RF signal above the LO signal frequency, at f'.sub.RF =(n)f.sub.LO +f.sub.IF, will also produce a signal at the monitored IF frequency. To resolve this ambiguity, the tunable bandpass filter 21 shown in FIG. 1 acts as a tunable bandpass filter over a frequency range including f.sub.RF, as shown by the broken line curve 31 that appears in FIG. 2, thereby attenuating any image signal 33 at f'.sub.RF. Therefore, the passband of the tunable bandpass filter 21 shown in FIG. 1 must track the sweeping LO signal, with the center frequency of the passband separated from the LO frequency (or harmonic) by the IF signal frequency.
The tunable bandpass filter 21 shown in FIG. 1 can be a YIG-tuned resonator filter, or preselector, which comprises a YIG sphere suspended between two orthogonal half loop conductors with the YIG sphere centered on the intersection of the loop axes. When the YIG sphere is not magnetized, an RF input signal is not transferred between the half loops, because there is no interaction between the RF input signal and the YIG sphere and the loops are perpendicular to each other. However, in the presence of an externally applied DC magnetic field along an axis perpendicular to the half loops, the dipoles in the YIG sphere align with the DC magnetic field, producing a strong net magnetization, M, in the YIG sphere. An RF input signal applied to the input half loop therefore produces an alternating RF magnetic field perpendicular to the externally applied DC magnetic field, which causes the dipoles in the YIG sphere to precess around the DC magnetic field at the frequency of the RF input signal. The precession frequency is equal to the frequency of the RF input signal if the RF input signal frequency equals or closely approximates the dipole resonance frequency of the spherical YIG resonator. The resonance frequency for a spherical YIG resonator is: EQU f.sub.r =.gamma.(H.sub.0 .+-.H.sub.a),
where H.sub.0 is the strength of the externally applied DC magnetic field in oersteds, H.sub.a is the internal anisotropy field (in oresteds) within the YIG material, and .gamma. is the gyromagnetic ratio (2.8 MHz/oersted).
If an RF input signal at or near f.sub.r is applied to the input half loop, the YIG material exhibits ferrimagnetic resonance, such that the precessing dipoles create a circularly polarized magnetic field, rotating at the RF input signal frequency, in a plane perpendicular to the externally applied DC magnetic field. This rotating magnetic field couples to the output half loop, including an RF signal in the output half loop that, at the resonance frequency f.sub.r, is phase-shifted 90.degree. from the RF input signal.
The YIG-tuned resonator filter therefore acts as a gyrator. The phase shift in one direction through the YIG-tuned resonator filter differs from the phase shift in the other direction by 180.degree.. The filtering function is achieved because RF input signals deviating from the dipole resonance frequency by more than a small amount do not couple to the YIG sphere.
Because the resonance bandwidth can be made fairly narrow, the YIG resonator comprises a highly selective bandpass filter at RF frequencies, tunable by varying the strength of the externally applied DC magnetic field. Typical loaded Q values for YIG-tuned resonator filters range from 100 to 400.
However, the prior art circuit shown in FIG. 1 has several drawbacks. RF input signals in the frequency range from 2.7 to 26.5 GHz are switched to the broadband YIG-tuned resonator filter 21 which is used as a preselector. A filtered signal is applied to the harmonic mixer 22, where it is mixed with the fundamental or harmonics of the swept, YIG-tuned local oscillator (LO) 23 to produce the predetermined IF output signal. This approach has the shortcoming that microwave harmonic mixers are inefficient, which drastically reduces the sensitivity of the microwave spectrum analyzer, especially when mixing with higher harmonics that are at high frequency. Fundamental mixing has been used to overcome the sensitivity degradation problem. This is achieved by means of a broadband LO (i.e., 5 to 26.5 GHz) or by multiplying the frequency of the signal from a narrowband LO (i.e., 3 to 6.8 GHz) to produce a broadband LO source for the harmonic mixer 22. While these approaches offer performance advantages, they are expensive to implement.
Additionally, a schematic diagram of one known integrated harmonic mixer 22 is shown in FIG. 1A. This mixer comprises a single diode D having a first end connected to the YIG-tuned resonator output coupling loop. The other end of the diode D is connected to one end of a transmission line TL whose other end is connected to an LO signal coupler. A narrowband, e.g., 3 to 6.8 GHz, LO signal is coupled via the LO signal coupler to the transmission line TL. RF, LO, and IF signals are not isolated. The RF, LO, and IF currents all flow through the transmission line TL. Harmonic mixing is accomplished by biasing the diode D through a resistor R. The following drawbacks are apparent.
The LO signal, RF signal, image, and multiple mixing products all must flow through the transmission line TL and are terminated through an AC load. No image enhancement, in which higher order mixing products are reflected back to the mixer for further mixing, occurs. Also, the AC load must be a broadband load to maintain the response of the mixer flat across the frequency range of interest. Furthermore, harmonic mixing is accomplished by biasing the diode D. This class of mixer (single diode biased for different harmonics) is not efficient.
A schematic diagram of another known harmonic mixer 22 is shown in FIG. 1B. This mixer is a single-balanced fundamental and odd order harmonic mixer, as described in U.S. Pat. No. 4,817,200. The LO and RF signals are isolated through the balun action of a full output coupling loop. This mixer is an odd harmonic mixer in which the IF signal at a frequency f.sub.IF =(2n+1)f.sub.LO .+-.f.sub.RF flows through port 1 and port 2. The LO and RF signals are isolated, eliminating a need for broadband microwave load needed in the case of single diode mixing in the mixer shown in FIG. 1A.
However, the IF signal must be diplexed from the LO signal by means of an inductor L and a capacitor C through port 1 and port 2. The inductor L is the return path for the IF signal at a frequency f.sub.IF =(2n+1)f.sub.LO -f.sub.RF, but mixing products at a frequency f.sub.IF =(2n+1)f.sub.LO +f.sub.RF flow through port 1 to the LO input. Therefore, mixing products at frequencies (2n+1)f.sub.LO +f.sub.RF (i.e., the image frequency) are transmitted through port 1 and must be terminated by the LO source impedance. No enhancement of images and multiples is achieved.